1.5 Hour Lamp Fader (Sunset Lamp)

1.5 Hour Lamp Fader (Sunset Lamp)

Similar to the one above, the sunset lamp comes on at full brightness and then slowly fades out over 1.5 hours time and stays off until power is recycled.

Automatic 12 Volt Lamp Fader

Automatic 12 Volt Lamp Fader

This circuit is similar to the "Fading Red Eyes" circuit (in the LED section) used to fade a pair of red LEDs. In this version, the lamps are faded by varying the duty cycle so that higher power incandescent lamps can be used without much power loss. The switching waveform is generated by comparing two linear ramps of different frequencies. The higher frequency ramp waveform (about 75 Hz.) is produced from one section of the LM324 quad op-amp wired as a Schmitt trigger oscillator. The lower frequency ramp controls the fading rate and is generated from the upper two op-amps similar to the "fading eyes" circuit. The two ramp waveforms at pins 9 and 1 are compared by the 4th op-amp which generates a varying duty cycle rectangular waveform to drive the output transistor. A second transistor is used to invert the waveform so that one group of lamps will fade as the other group brightens. The 2N3053 will handle up to 500 milliamps so you could connect 12 strings of 4 LEDs each (48 LEDs) with a 220 ohm resistor in series with each group of 4 LEDs. This would total about 250 milliamps. Or you can use three 4 volt, 200 mA Xmas tree bulbs in series. For higher power 12 volt automobile lamps, the transistor will need to be replaced with a MOSFET that can handle several amps of current. See the drawing below the schematic for possible hookups.

Other possible hookups

12 Volt Lamp Dimmer

12 Volt Lamp Dimmer

Here is a 12 volt / 2 amp lamp dimmer that can be used to dim a standard 25 watt automobile brake or backup bulb by controlling the duty cycle of a astable 555 timer oscillator. When the wiper of the potentiometer is at the uppermost position, the capacitor will charge quickly through both 1K resistors and the diode, producing a short positive interval and long negative interval which dims the lamp to near darkness. When the potentiometer wiper is at the lowermost position, the capacitor will charge through both 1K resistors and the 50K potentiometer and discharge through the lower 1K resistor, producing a long positive interval and short negative interval which brightens the lamp to near full intensity. The duty cycle of the 200 Hz square wave can be varied from approximately 5% to 95%. The two circuits below illustrate connecting the lamp to either the positive or negative side of the supply.

FM Beacon Broadcast Transmitter (88-108 MHz)

FM Beacon Broadcast Transmitter (88-108 MHz)

This circuit will transmit a continuous audio tone on the FM broadcast band (88-108 MHz) which could used for remote control or security purposes. Circuit draws about 30 mA from a 6-9 volt battery and can be received to about 100 yards. A 555 timer is used to produce the tone (about 600 Hz) which frequency modulates a Hartley oscillator. A second JFET transistor buffer stage is used to isolate the oscillator from the antenna so that the antenna position and length has less effect on the frequency. Fine frequency adjustment can be made by adjusting the 200 ohm resistor in series with the battery. Oscillator frequency is set by a 5 turn tapped inductor and 13 pF capacitor. The inductor was wound around a #8 X 32 bolt (about 3/16 diameter) and then removed by unscrewing the bolt. The inductor was then streached to about a 3/8 inch length and tapped near the center. The oscillator frequency should come out somewhere near the center of the band (98 MHz) and can be shifted higher or lower by slightly expanding or compressing the inductor. A small signal diode (1N914 or 1N4148) is used as a varactor diode so that the total capacity in parallel with the inductor varies slightly at the audio rate thus causing the oscillator frequency to change at the audio rate (600 Hz). The ramping waveform at pins 2 and 6 of the timer is applied to the reversed biased diode through a large (1 Meg) resistor so that the capacitance of the diode changes as the ramping voltage changes thus altering the frequency of the tank circuit. Alternately, an audio signal could be applied to the 1 Meg resistor to modulate the oscillator but it may require an additional pullup resistor to reverse bias the diode. The N channel JFET transistors used should be high frequency VHF or UHF types (Radio Shack #276-2062 MPF102) or similar.

Micro Power AM Broadcast Transmitter

Micro Power AM Broadcast Transmitter

In this circuit, a 74HC14 hex Schmitt trigger inverter is used as a square wave oscillator to drive a small signal transistor in a class C amplifier configuration. The oscillator frequency can be either fixed by a crystal or made adjustable (VFO) with a capacitor/resistor combination. A 100pF capacitor is used in place of the crystal for VFO operation. Amplitude modulation is accomplished with a second transistor that controls the DC voltage to the output stage. The modulator stage is biased so that half the supply voltage or 6 volts is applied to the output stage with no modulation. The output stage is tuned and matched to the antenna with a standard variable 30-365 pF capacitor. Approximately 20 milliamps of current will flow in the antenna lead (at frequencies near the top of the band) when the output stage is optimally tuned to the oscillator frequency. A small 'grain of wheat' lamp is used to indicate antenna current and optimum settings. The 140 uH inductor was made using a 2 inch length of 7/8 inch (OD) PVC pipe wound with 120 turns of #28 copper wire. Best performance is obtained near the high end of the broadcast band (1.6 MHz) since the antenna length is only a very small fraction of a wavelength. Input power to the amplifier is less than 100 milliwatts and antenna length is 3 meters or less which complies with FCC rules. Output power is somewhere in the 40 microwatt range and the signal can be heard approximately 80 feet. Radiated power output can be approximated by working out the antenna radiation resistance and multiplying by the antenna current squared. The radiation resistance for a dipole antenna less than 1/4 wavelength is

R = 80*[(pi)^2]*[(Length/wavelength)^2]*(a factor depending on the form of the current distribution) The factor depending on the current distribution turns out to be [(average current along the rod)/(feed current)]^2 for short rods, which is 1/4 for a linearly-tapered current distribution falling to zero at the ends. Even if the rods are capped with plates, this factor cannot be larger than 1. Substituting values for a 9.8 foot dipole at a frequency of 1.6 MHz we get R= 790*.000354*.25 = .07 Ohms. And the resistance will be only half as much for a monopole or 0.035 Ohms. Radiated power at 20 milliamps works out to about I^2 * R = 14 microwatts.

Parallel Port Relay Interface

Parallel Port Relay Interface

Below are three examples of controlling a relay from the PC's parallel printer port (LPT1 or LPT2). Figure A shows a solid state relay controlled by one of the parallel port data lines (D0-D7) using a 300 ohm resistor and 5 volt power source. The solid state relay will energize when a "0" is written to the data line. Figure B and C show mechanical relays controlled by two transistors. The relay in figure B is energized when a "1" is written to the data line and the relay in figure C is energized by writing a "0" to the line. In each of the three circuits, a common connection is made from the negative side of the power supply to one of the port ground pins (18-25).

There are three possible base addresses for the parallel port You may need to try all three base addresses to determine the correct address for the port you are using but LPT1 is usually at Hex 0378. The QBasic "OUT" command can be used to send data to the port. OUT, &H0378,0 sets D0-D7 low and OUT, &H378,255 sets D0-D7 high. The parallel port also provides four control lines (C0,C1,C2,C3) that can be set high or low by writing data to the base address+2 so if the base address is Hex 0378 then the address of the control latch would be Hex 037A. Note that three of the control bits are inverted so writing a "0" to the control latch will set C0,C1,C3 high and C2 low.

Simple Op-Amp Radio

Simple Op-Amp Radio

This is basically a crystal radio with an audio amplifier which is fairly sensitive and receives several strong stations in the Los Angeles area with a minimal 15 foot antenna. Longer antennas will provide a stronger signal but the selectivity will be worse and strong stations may be heard in the background of weaker ones. Using a long wire antenna, the selectivity can be improved by connecting it to one of the taps on the coil instead of the junction of the capacitor and coil. Some connection to ground is required but I found that standing outside on a concrete slab and just allowing the long headphone leads to lay on the concrete was sufficient to listen to the local news station (KNX 1070). The inductor was wound with 200 turns of #28 enameled copper wire on a 7/8 diameter, 4 inch length of PVC pipe, which yields about 220 uH. The inductor was wound with taps every 20 turns so the diode and antenna connections could be selected for best results which turned out to be 60 turns from the antenna end for the diode. The diode should be a germanium (1N34A type) for best results, but silicon diodes will also work if the signal is strong enough. The carrier frequency is removed from the rectified signal at the cathode of the diode by the 300 pF cap and the audio frequency is passed by the 0.1uF capacitor to the non-inverting input of the first op-amp which functions as a high impedance buffer stage. The second op-amp stage increases the voltage level about 50 times and is DC coupled to the first through the 10K resistor. If the pairs of 100K and 1 Meg resistors are not close in value (1%) you may need to either use closer matched values or add a capacitor in series with the 10K resistor to keep the DC voltage at the transistor emitter between 3 and 6 volts. Another approach would be to reduce the overall gain with a smaller feedback resistor (470K). High impedance headphones will probably work best, but walkman stereo type headphones will also work. Circuit draws about 10 mA from a 9 volt source. Germanium diodes (1N34A) types are available from Radio Shack, #276-1123.

Thermostat for 1KW Space Heater (SCR controlled)

Thermostat for 1KW Space Heater (SCR controlled)

Below is a thermostat circuit I recently built to control a 1300 watt space heater. The heater element (not shown) is connected in series with two back to back 16 amp SCRs (not shown) which are controlled with a small pulse transformer. The pulse transformer has 3 identical windings, two of which are used to supply trigger pulses to the SCRs, and the third winding is connected to a PNP transistor pair that alternately supply pulses to the transformer at the beginning of each AC half cycle. The trigger pulses are applied to both SCRs near the beginning of each AC half cycle but only one conducts depending on the AC polarity.

DC power for the circuit is shown in the lower left section of the drawing and uses a 1.25uF, 400 volt non-polarized capacitor to obtain about 50mA of current from the AC line. The current is rectified by 2 diodes and used to charge a couple larger low voltage capacitors (3300uF) which provide about 6 volts DC for the circuit. The DC voltage is regulated by the 6.2 volt zener and the 150 ohm resistor in series with the line limits the surge current when power is first applied.

The lower comparator (output at pin 13) serves as a zero crossing detector and produces a 60 Hz square wave in phase with the AC line. The phase is shifted slightly by the 0.33 uF, 220K and 1K network so that the SCR trigger pulse arrives when the line voltage is a few volts above or below zero. The SCRs will not trigger at exactly zero since there will be no voltage to maintain conduction.

The upper two comparators operate in same manner as described in the "Electronic thermostat and relay" circuit. A low level at pin 2 is produced when the temperature is above the desired level and inhibits the square wave at pin 13 and prevents triggering of the SCRs. When the temperature drops below the desired level, pin 2 will move to an open circuit condition allowing the square wave at pin 13 to trigger the SCRs.

The comparator near the center of the drawing (pins 8,9,14) is used to allow the heater to be manually run for a few minutes and automatically shut off. A momentary toggle switch (shown connected to a 51 ohm resistor) is used to discharge the 1000uF capacitor so that pin 2 of the upper comparator moves to a open circuit state allowing the 60 Hz square wave to trigger the SCRs and power the heater. When the capacitor reaches about 4 volts the circuit returns to normal operation where the thermistor controls the operation. The momentary switch can also be toggled so that the capacitor charges above 4 volts and shuts off the heater if the temperature is above the setting of the pot.

Electronic Thermostat and Relay Circuit

Electronic Thermostat and Relay Circuit

Here is a simple thermostat circuit that can be used to control a relay and supply power to a small space heater through the relay contacts. The relay contacts should be rated above the current requirements for the heater.

Temperature changes are detected by a (1.7K @ 70F) thermistor placed in series with a 5K potentiometer which produces about 50 millivolts per degree F at the input of the LM339 voltage comparator. The two 1K resistors connected to pin 7 set the reference voltage at half the supply voltage and the hysteresis range to about 3 degrees or 150 millivolts. The hysteresis range (temperature range where the relay engages and disengages) can be adjusted with the 10K resistor between pins 1 and 7. A higher value will narrow the range.

In operation, the series resistor is adjusted so that the relay just toggles off at the desired temperature. A three degree drop in temperature should cause the relay to toggle back on and remain on until the temperature again rises to the preset level. The relay action can be reversed so it toggles off at the lower end of the range by reversing the locations of the 5K potentiometer and thermistor. The 5.1 volt zener diode regulates the circuit voltage so that small changes in the 12 volt supply will not effect operation. The voltage across the thermistor should be half the supply or about 2.6 volts when the temperature is within the 3 degree range set by the potentiometer. Most any thermistor can be used, but the resistance should be above 1K ohm at the temperature of interest. The series resistor selected should be about twice the resistance of the thermistor so the adjustment ends up near the center of the control.

9 Second Digital Readout Countdown Timer

This circuit provides a visual 9 second delay using a 7 segment digital readout LED. When the switch is closed, the CD4010 up/down counter is preset to 9 and the 555 timer is disabled with the output held high. When the switch is opened, the timer produces an approximate 1 second clock signal, decrementing the counter until the 0 count is reached. When the zero count is reached, the 'carry out' signal at pin 7 of the counter moves low, energizing the 12 volt relay and stopping the clock with a low signal on the reset line (pin 4). The relay will remain energized until the switch is again closed, resetting the counter to 9. The 1 second clock signal from the 555 timer can be adjusted slightly longer or shorter by increasing or decreasing the resistor value at pin 3 of the timer.

The CD4510 is a CMOS Presettable BCD Up/Down counter which can be preset to any number between 0 and 9 with a high level on the PRESET ENABLE line, (pin 1) or reset to 0 with a high level on the RESET line (pin 9). Inputs for presetting the counter (P1, P2, P3, P4) are on pins (4, 12, 13, 3) respectively. The counter advances up or down on each positive-going clock transition (pin 15) and the counting direction (up or down) is controlled by the logic level on the UP/DOWN input (pin 10, high=up, low=down). The CARRY-IN signal (pin 5) disables the counter with a high logic level.

The CD4511 is a CMOS BCD to 7 segment latch decoder capable of sourcing up to 25 mA which allows it to drive LEDs and other displays directly. A LATCH-ENABLE line (pin 5, active low) stores data from the BCD input lines. A LAMP-TEST input (pin 3, active low) can be used to illuminate all 7 segments, and a BLANKING input (pin 4, active low) can be used to turn all segments off. The LED display must be a common cathode type so that the segments are illuminated with a positive voltage on their respective connections. Complete data sheets for the CD4510 and CD4511 can be obtained by answer fax from

9 Second LED Timer and Relay Circuit

9 Second LED Timer and Relay Circuit

This circuit provides a visual 9 second delay using 10 LEDs before closing a 12 volt relay. When the reset switch is closed, the 4017 decade counter will be reset to the 0 count which illuminates the LED driven from pin 3. The 555 timer output at pin 3 will be high and the voltage at pins 6 and 2 of the timer will be a little less than the lower trigger point, or about 3 volts. When the switch is opened, the transistor in parallel with the timing capacitor (22uF) is shut off allowing the capacitor to begin charging and the 555 timer circuit to produce an approximate 1 second clock signal to the decade counter. The counter advances on each positive going change at pin 14 and is enabled with pin 13 terminated low. When the 9th count is reached, pin 11 and 13 will be high, stopping the counter and energizing the relay. Longer delay times can be obtained with a larger capacitor or larger resistor at pins 2 and 6 of the 555 timer.

Power-Off Time Delay Relay

Power-Off Time Delay Relay

The two circuits below illustrate opening a relay contact a short time after the ignition or ligh switch is turned off. The capacitor is charged and the relay is closed when the voltage at the diode anode rises to +12 volts. The circuit on the left is a common collector or emitter follower and has the advantage of one less part since a resistor is not needed in series with the transistor base. However the voltage across the relay coil will be two diode drops less than the supply voltage, or about 11 volts for a 12.5 volt input. The common emitter configuration on the right offers the advantage of the full supply voltage across the load for most of the delay time, which makes the relay pull-in and drop-out voltages less of a concern but requires an extra resistor in series with transistor base. The common emitter (circuit on the right) is the better circuit since the series base resistor can be selected to obtain the desired delay time whereas the capacitor must be selected for the common collector (or an additional resistor used in parallel with the capacitor). The time delay for the common emitter will be approximately 3 time constants or 3*R*C. The capacitor/resistor values can be worked out from the relay coil current and transistor gain. For example a 120 ohm relay coil will draw 100 mA at 12 volts and assuming a transistor gain of 30, the base current will be 100/30 = 3 mA. The voltage across the resistor will be the supply voltage minus two diode drops or 12-1.4 = 10.6. The resistor value will be the voltage/current = 10.6/0.003 = 3533 or about 3.6K. The capacitor value for a 15 second delay will be 15/3R = 1327 uF. We can use a standard 1000 uF capacitor and increase the resistor proportionally to get 15 seconds.

Power-On Time Delay Relay

Power-On Time Delay Relay

Here's a power-on time delay relay circuit that takes advantage of the emitter/base breakdown voltage of an ordinary bi-polar transistor. The reverse connected emitter/base junction of a 2N3904 transistor is used as an 8 volt zener diode which creates a higher turn-on voltage for the Darlington connected transistor pair. Most any bi-polar transistor may be used, but the zener voltage will vary from about 6 to 9 volts depending on the particular transistor used. Time delay is roughly 7 seconds using a 47K resistor and 100uF capacitor and can be reduced by reducing the R or C values. Longer delays can be obtained with a larger capacitor, the timing resistor probably shouldn't be increased past 47K. The circuit should work with most any 12 volt DC relay that has a coil resistance of 75 ohms or more. The 10K resistor connected across the supply provides a discharge path for the capacitor when power is turned off and is not needed if the power supply already has a bleeder resistor.

Low Voltage, High Current Time Delay Circuit

Low Voltage, High Current Time Delay Circuit

In this circuit a LM339 quad voltage comparator is used to generate a time delay and control a high current output at low voltage. Approximately 5 amps of current can be obtained using a couple fresh alkaline D batteries. Three of the comparators are wired in parallel to drive a medium power PNP transistor (2N2905 or similar) which in turn drives a high current NPN transistor (TIP35 or similar). The 4th comparator is used to generate a time delay after the normally closed switch is opened. Two resistors (36K and 62K) are used as a voltage divider which applies about two-thirds of the battery voltage to the (+) comparator input, or about 2 volts. The delay time after the switch is opened will be around one time constant using a 50uF capacitor and 100K variable resistor, or about (50u * 100K) = 5 seconds. The time can be reduced by adjusting the resistor to a lower value or using a smaller capacitor. Longer times can be obtained with a larger resistor or capacitor. To operate the circuit on higher voltages, the 10 ohm resistor should be increased proportionally, (4.5 volts = 15 ohms).

Capacitor Discharge Ignition Circuit (CDI)

Capacitor Discharge Ignition Circuit (CDI)

The CDI ignition circuit produces a spark from an ignition coil by discharging a capacitor across the primary of the coil. A 2uF capacitor is charged to about 340 volts and the discharge is controlled by an SCR. A Schmitt trigger oscillator (74C14) and MOSFET (IRF510) are used to drive the low voltage side of a small (120/12 volt) power transformer and a voltage doubler arrangement is used on the high voltage side to increase the capacitor voltage to about 340 volts. A similar Schmitt trigger oscillator is used to trigger the SCR about 4 times per second. The power supply is gated off during the discharge time so that the SCR will stop conducting and return to it's blocking state. The diode connected from the 3904 to pin 9 of the 74C14 causes the power supply oscillator to stop during discharge time. The circuit draws only about 200 milliamps from a 12 volt source and delivers almost twice the normal energy of a conventional ignition circuit. High voltage from the coil is about 10KV using a 3/8 inch spark gap at normal air temperature and pressure. Spark rate can be increased to possibly 10 Hertz without losing much spark intensity, but is limited by the low frequency power transformer and duty cycle of the oscillator. For faster spark rates, a higher frequency and lower impedance supply would be required. Note that the ignition coil is not grounded and presents a shock hazard on all of it's terminals. Use CAUTION when operating the circuit. An alternate method of connecting the coil is to ground the (-) terminal and relocate the capacitor between the cathode of the rectifier diode and the positive coil terminal. The SCR is then placed between ground and the +340 volt side of the capacitor. This reduces the shock hazard and is the usual configuration in automotive applications.

Long Loopstick Antenna

Long Loopstick Antenna

Wound on a 3 foot length of PVC pipe, the long loopstick antenna was an experiment to try to improve AM radio reception without using a long wire or ground. It works fairly well and greatly improved reception of a weak station 130 miles away. A longer rod antenna will probably work better if space allows. The number of turns of wire needed for the loopstick can be worked out from the single layer, air core inductance formula:

Inductance = (radius^2 * turns^2) / ((9*radius)+(10*length))

where dimensions are in inches and inductance is in microhenrys. The inductance should be about 230 microhenrys to operate with a standard AM radio tuning capacitor (33-330 pF). The 3 foot PVC pipe is wound with approximately 500 evenly spaced turns of #24 copper wire which forms an inductor of about 170 microhenrys, but I ended up with a little more (213uH) because the winding spacing wasn't exactly even. A secondary coil of about 50 turns is wound along the length of the pipe on top of the primary and then connected to 4 turns of wire wound directly around the radio. The windings around the radio are orientated so that the radio's internal antenna rod passes through the external windings. A better method of coupling would be to wind a few turns directly around the internal rod antenna inside the radio itself, but you would have to open the radio to do that. In operation, the antenna should be horizontal to the ground and at right angles to the direction of the radio station of interest. Tune the radio to a weak station so you can hear a definite amount of noise, and then tune the antenna capacitor and rotate the antenna for the best response. The antenna should also be located away from lamp dimmers, computer monitors and other devices that cause electrical interference.

Delayed Turn-On Relay

Parts List:
R1,R3 = 10K              Q1 = 2N3906, or equivalent
R2 = 680K (see text)     IC1 = 4001, or equivalent
R4,R5 = 6K8              D1,D2,D3 = 1N4001, or equivalent
C1 = see text            Ry = Relay, 12V
C2 = 0.1µF, ceramic

This circuit is a delayed turn-on relay driver and can produce time delays for up to several minutes with reasonable accuracy.

The 14001 (or 4001) CMOS gate here is configured as a simple digital inverter. Its output is fed to the base of a regular 2N3906 (PNP) transistor, Q1, at the junction of resistor R5 and capacitor C2. The input to IC1 is taken from the junction of the time-controlled potential divider formed by R2 and C1. Before power is applied to the circuit, C1 is fully discharged. Therefore, the inverter input is grounded, and its output equals the positive supply rail; Q1 and RY1 are both off under this circuit condition. When power is applied to the circuit, C1 charges through R2, and the exponentially rising voltage is applied to the input of the CMOS inverter gate.

After a time delay determined by the RC time constant values of C1 and R2, this voltage rises to the threshold value of the CMOS inverter gate. The gate's output then falls toward zero volts and drives Q1 and relay RY 'ON'. The relay then remains on until power is removed from the circuit. When that occurs, capacitor C1 discharges rapidly through diode D1 and R1, completing the sequence.

The time delay can be controlled by different values for C1 and R2. The delay is approximately 0.5 seconds for every µF as value for C1. The delay can further be made variable by replacing R2 with a fixed and a variable resistor equal to that of the value of R2. Taken the value for R2 of 680K, it would be a combination of 180K for the fixed resistor in series with a 500K variable trim pot. The fixed resistor is necessary.

Passive Aircraft Receiver

Passive Aircraft Receiver

The Passive Aircraft Receiver is basically an amplified "crystal radio" designed to receive nearby AM aircraft transmissions. The "passive" design uses no oscillators or other RF circuitry capable of interfering with aircraft communications so it should be fine inside the cabin of the aircraft. Nevertheless, check the regulations before using this receiver on a commercial airliner. New security regulations probably prohibit this device on commercial flights. Do not expect to hear two-way aircraft transmissions with this receiver! It is a short-range receiver only.

The detector diode is a 1N5711, HP2835 or similar Schottky detector diode. The 10 megohm resistors provide a small diode bias current for better detector efficiency. The tuning capacitor may be any small variable with a range from about 5 pF to about 15 or 20 pF. The 0.15 uH inductor may be a molded choke or a few turns wound with a small diameter. Experiment with the coil to get the desired tuning range. The aircraft frequencies are directly above the FM band so a proper inductor will tune FM stations with the capacitor set near maximum capacity. (The FM stations will sound distorted since they are being slope detected.) Other capacitor and inductor combinations may be selected to tune other bands if desired. (Try the CB band at 27 MHz.) The LM358 dual op-amp draws under 1 ma so the battery life is quite long. A speaker amplifier may be added to drive a speaker or low-z earphone. The antenna can be a couple of inches if the receiver is near the transmitter or a couple of feet for maximum range. The selectivity is reduce as the antenna length is increased so best performance is achieved with the shortest acceptable antenna. Try increasing the 1.8 pF capacitor value when using very short antennas and decreasing it for long antennas. The receiver could be built into a small plastic box with a short antenna inside.

40 m Band Direct Conversion Receiver

40 m Band Direct Conversion Receiver

Building a practical and usable direct conversion receiver for the 40 m CW band is not as simple as it might appear. Broadcast station signals from the adjacent 41 m band, will easily overload most direct conversion mixer designs with their unwanted (and quite nearby) S9 +40 dB signals. My solution incorporates a diode-ring mixer and a narrow band rf input filter. These choices result in observed overall receiver dynamic range than that experienced when using an IC mixer module (such as a NE612). Comfortable and undisturbed operation in the evening was possible when tested using my Windom antenna. Some of this project's objectives included:

• High dynamic range diode-ring mixer
• Stable oscillator with a 30 kHz tuning range
• Tuning range from 7005 kHz to 7035 kHz
• Narrow front end band pass input filter
• Broadband 50 Ohm termination
• Audio selectivity used in the AF amplifier
• Symmetrical (differential coupling) design
• Low battery / power supply current consumption
• 60 Ohm headphone output impedance

You may recognize many of the stage circuits - as they are similar to those used in some of my other projects. The VFO and the RF input band pass filter are each designed around ceramic resonators. These are becoming difficult to locate. [See the note below.] I have used a HPF505 type ring diode mixer - but other mixers such as the IE500 or SRA1 should be suitable. The diode ring mixer output drives a parallel arrangement consisting of R11 and the differential input impedances of IC1. This results in a more stable 50-Ohm broadband termination for the mixer output (by contrast to that possible on a typical RCL based diplexer).

The gain of the broadband amplifier following the mixer, IC1, is set by the choice of R10. A 40 dB gain is achieved by using 100-Ohm resistor at R10. Any IC1 rf output signals are shunted to ground - leaving only an audio signal to be applied to the following stage. IC2 is an operational amplifier that is used to amplify the remaining audio signals by yet an additional 46 dB. Passive audio filter components are used at the input of IC2 so that only signals at or near 750 Hz are amplified. The overall gain and output level is almost too great for comfortable headphone listening. An rf attenuator (P1 located near the antenna input terminal) is used as a means of controlling the receiver output volume

Perhaps the most critical objective in this design is the voltage present at Pin 6 of IC1 (broadband amplifier). Adjust the value of R7 slightly as needed to set the voltage at IC1 pin 6 to be very close to +8 V. IC1 (at its output pins 4 and 5) provides the dc bias voltage of +6 V (Ub/2) as required for the non-inverting opamp inputs of the final amplifier stage, IC2. The dc input bias for this circuit is relatively critical if distortion is to be minimized. This opamp circuit can provide an undistorted audio output (at pins 1 and 7 of IC2) of as much as 10 Vss. When properly built and adjusted, this receiver should consume only about 25 mA with the antenna disconnected. The receiver functions well at any power supply voltage ranging from +15 V down to as low as +9 V.

Parts No.	Value
R1	27 kOhm
R2	22 kOhm
R3	470 Ohm
R4	330 Ohm
R5,6	4,7 kOhm
R7	220 Ohm
R8,9	10 kOhm
R10	100 Ohm
R11	51 Ohm
R12,13	1 kOhm
P1	1 kOhm, linear
C1	20 ... 325 pF, variable capacitor
C2	100 pF
C3	680 pF
C4	330 pF
C5,11,12	4,7 nF
C6,7	0,1 µF
C8,9	47 nF
C10	2,2 uF, no electrolytice cap
C13	100 uF, 25 V electrolytic cap
Dr1	33 mH
L1,2	FT37-43, 5 turns tap at the second turn
Q1,2	SFE 7.02 M2C, Murata
VT1	2N3904
IC1	NE592-N8 DIP
IC2	NE5532 DIP, dual opamp
M	Mixer HPF505, IE500, SRA1 e.g.
KH	Headphones Ri > 60 Ohm ( 32 + 32 Ohm)

33 Volt DC to DC Converter

33 Volt DC To DC Converter

Description: 33 Volt DC To DC Converter

This is the right solution for your MOSFET based low power linear amplifier if you have no AC source in your house and have to rely on battery. Generally IRF MOSFET transistors require DC supply at least 24 to 36 volts for efficient operation and their output power drops drastically when the supply voltage is only about 12 volts. The dc to dc converter described here is very simple and enables you to apply 33 volts to IFR transistors from your 12 volt rechargeable battery. Another advantage is that you can reduce the size of your power transformer if you are working on mains. It will also come handy for field days when you have to depend on battery for operating your rig without mains supply. A pair of 2N3055 , work as an oscillator at high frequency and the higher voltage appearing across the secondary of the oscillator coil rectified and filtered to get 33 volts dc. The oscillator coil is wound on 30 mm diameter toroid, which is available easily anywhere .Two of them are used after fixing them together with quickfix. The primary containing 9+9 turns is wound with 20 awg enameled copper wire and the feedback winding 4+4 turns is wound over it. The secondary is wound using 22 awg wire, about 36 turns. This power supply should be switched off, when receiving as otherwise , it may produce noise in receiver. Four numbers of high speed switching diodes BA 157 and 1000 mF/ 50 volts filter capacitor produces 33 volt DC. The usual rectifiers diodes are not suitable here because, the voltage is at high frequency. The two transistors [T1 and T2 ] are mounted on a heat sink it is found that the transistors get only slight warm during transmission.

30 Meter Receiver Project

30 Meter Receiver Project

Background

A lot can be learned when using strict design criteria to build a project. I set out to build an entire receiver using only 2N3904 transistors and at the end settled upon the design shown above. This design resembles that of the Ugly Direct receiver on this web site in a lot of ways and is also a low-cost popcorn project. A great deal of time was spent building and testing various VFO designs and investigating an interesting single-balanced mixer using two 2N3904 BJT's. The design process and reasons for abandoning my original criteria in the case of the mixer and VFO will be discussed.

Bandpass Filter

A bandpass filter was designed for low insertion loss to help maintain the receiver noise figure. In keeping with this, NP0 ceramic capacitors were used for the 68 pF and 5 pF fixed-value capacitors. The trimmer cap was a 5 -20 pF ceramic variable with a Qu of 300. (DigiKey bottom-adjusted SG20016-ND). The leads were bent so that each trimmer cap could be adjusted from the top.

The L1 and L2 inductors were wound using 27 turns of #26 AWG enamel coated wire on T50-6 powdered iron toroids. A tap was made four turns up from the grounded end. Qu is ~ 250 for these inductors. The center frequency is 10.125 MHz, the bandwidth is 0.88 MHz and the loaded Q of the resonators is 11.5.

The easiest method to tune the resonators is to peak the trimmer caps for the greatest measured output voltage using an oscilloscope. I used the receiver VFO temporarily terminated with a -10dB, 50 ohm pad to obtain the correct filter input impedance and connected it to the input end of the filter. I temporarily terminated the output of the filter with a 51 ohm resistor to ground. The VFO was tuned to the center frequency by placing it next to a receiver set on 10.125 MHz. A frequency counter can also be used. The trimmers were adjusted on each resonator to obtain the highest measured voltage possible. The filter was then placed in the receiver after removing the temporary alterations used during calibration.

If you do not have access to test equipment, tune the resonators at the center frequency while listening to the receiver in the headphones to obtain the greatest possible band noise. Confirm your adjustments by tweaking the trim caps while listening to a QSO as well.

Product Detector

A product detector using either one or more 2N3904 transistors was originally planned and indeed, four designs were built and tested. The 2 favorite detectors were a single-ended detector built with a single BJT which maybe used in an future novelty transceiver project and a passive mixer invented by Dr. Ulrich Rohde. The original mixer called for 2N5179 transistors and used a 0.1 uF coupling cap to the diplexer stage for RF output. It should have a VCC of 9 volts DC.

The mixer as built for this project is shown below.

The mixer as designed by Rohde had a reported IP3 of 33 dBm with a LO drive of 15-17 dBm and an insertion loss of ~ 6dB. This mixer operates in push-pull and the 22 ohm resistors on the transistor emitters provide degenerative feedback which makes component matching unnecessary. The schematic and brief write up can be found in QST for June 1994 in an article entitled Key Components of Modern Receiver Design-Part 2.

I built 2 versions of Rohde's mixer and tested them both in the receiver shown in the main schematic. I later discarded this design and replaced it with the familar diode ring mixer for the following subjective reasons; I noted a greater insertion loss, more hum and noise, higher LO drive level requirements and more WWV AM interference when compared to a diode ring mixer.

No quantitative measurements of the mixer were made. Listening tests and observations were only performed. Careful shielding of one version of the mixer resulted in a major improvement in hum and obliteration of an audio feedback problem noted when the AF gain was increased maximally when compared to the unshielded second version of the mixer. In addition, better performance would most certainly be realized if 2N5179 BJT's had been used instead of 2N3904's. Rohde's mixer certainly warrants further and better analysis with quantitative testing for use in home built receivers. If you build and test this mixer, please forward or publish the results for use by the Amateur Radio community. The trifilar wound transformers are identical to those shown elsewhere on this site and have phasing dots and coil numbering included for reference. Ugly constructing this mixer is extremely easy to do.

The diode ring mixer ultimately used has 50 ohm ports and can be a homebrew or commercial unit such as the popular SBL-1 from MiniCircuits.

VFO Design

Reviewing the Amateur Radio literatue revealed that JFETS enjoy tremendous popularity as the active device in LC local oscillators during the past ten years. To conform to the original design criteria of this project it was decided to build the VFO from only 2N3904s for the oscillator and the buffer sections. Four different VFO's were built and tested for short and long term frequency stability. Two partial schematics are shown below. Each design used the same buffer/amplifier for some sort of control. I found that it is possible to build very stable oscillators using the 2N3904, providing good quality, temperature-stable components are used. Careful attention to the design guidelines published by people like W1FB, W7EL and W7ZOI are mandatory. Electrical engineering knowledge would also be very helpful as I found biasing and feedback resistance values, coupling cap values and inductor Q all can have an effect on frequency stability and output noise.

My tests failed to determine why the JFET is so popular; there are just too many variables to factor in both electronically and through building techniques. Possibly, the easiest no-fail VFO to build is the tapped inductor Hartley using a JFET and this may help explain the popularity of the JFET. A JFET is probably a better choice with regard to phase noise because of a generally good noise figure and extremely low flicker noise. Despite the fact that the oscillators built with the bipolar transistors were very stable, one VFO stood out and was used. I have it displayed on this website as the project entitled:

An (LC) VFO for 30 Meters
This design was by far the most stable design for both short and long term drift and is the most stable VFO that I have ever built.

The VFO will see duty as a lab oscillator for use in future projects built for the great QRP band, 30 meters.

Diplexer

Presented is a Roy Lewellyn, W7EL diplexer design which provides a 50 ohm termination for the product detector at all frequencies. This single-pole filter has a 3dB cutoff design for 5.6 KHz. This diplexer design is used by permission. The 1.4 millihenry inductor is easily wound using a single layer on a FT50-77 ferrite toroid. Wind 38 turns of #26 AWG enamel coated wire with close spacing. If the builder only has access to the more common FT37-43 ferrite core, a 1.4 mH inductor can be wound using a 26 inch piece of #30 AWG wire. To construct this inductor, cut the 30 guage wire exactly 26 inches long and place one end of the piece of wire one inch through the ferrite toroid core. Begin wrapping the core with the other end of the wire in the usual fashion, proceeding carefully around the core avoiding knots and tangles. When you reach the original end of the wire continue winding past it and proceed around the core until you have a one inch length remaining. The second winding only partially covers the core. Use fairly tight loops on each winding to avoid getting a low inductance. The one inch leads should be ample for connecting to the circuit.

The wound inductor should be cemented face down onto the PC board after removing a small portion of copper big enough to fit the inductor on so that it is not touching any of the PCB copper surface. I used a hobby tool and sanded off the copper in a circular shape about 3/4 inch in diameter. The inductor was glued on with epoxy. The Qu of these home spun audio inductors is very low and consequently have very low loss. The 0.56uF cap I used was a miniaturized metallized polyester film (DigiKey EF2564-ND) which is an expensive part at 95 cents Canadian currency.

AF Preamp Chain

Following the diplexer is the familiar grounded base amplifier popularized by Roy Lewellyn, W7EL. This stage presents a low noise, wideband ~50 ohm input impedance to the diode ring detector and diplexer. An active decoupler is used to help prevent any hum getting into this stage. The 22uF capacitor in the decoupler circuit is capacitively multiplied by the beta of Q1 and has an effective filtering value of 22000 uF. The second stage is an amp designed by Wes Hayward, W7ZOI. The DC negative feedback provides bias stabalization for this stage. It is interesting to note that W7ZOI made a break in the DC feedback loop with a 10uF cap to ground so that there is no negative AC feedback around the amplifier and it operates at maximum gain.

Lowpass Filters

The source follower and two low pass stages were pulled from Solid State Design for The Radio Amateur published by the American Radio Relay League. The original article had the a ~1KHz cutoff frequency using 3K3 ohm resistors. The above schematic uses two 3K9 ohm resistors in each low pass stage for a cutoff frequency of 870 Hz. Other cutoff frequencies can be set by adjusting these resistor values as desired. The lowpass filter stages serve to improve QRM copy ability and attenuate a lot of the wideband noise generated and/or boosted in the preceeding stages.

AF Amp and Driver

Driving the final amp is a high gain common-emitter amp with its output connected to a 10K pot for volume control. The 0.0022 uF bypass cap is used as a highpass filter to help remove hiss.The final AF amp is a simple common-collector amp set for approximately 37 mA of emitter current. The 180 ohm resistor could be dropped to 150 ohm (~45 mA Ie) providing a heat sink is used on the BJT. A piece of PC board glued to the flat part of the transistor could be used to fashion a heat sink if you decide to stand more current than the original design.The 10 ohm resistor and the 22uF capacitor on the collector of Q8 form an RC filter to decouple the AF stage from the positive voltage supply. I have found this amp sufficient to drive a pair of Walkman style headphones with reasonable volume. Do not expect ear-shattering volumes levels however. Three sets of cheap headphones were tried and one pair gave very low volume when compared to the other sets. Keep this in mind if your not getting reasonable volume to your ears. The headphone jack used for this rig is a 1/8 inch (3.5 mm) stereo jack with both channels connected together for monoaural output.

Construction Hints

Like all electronic projects, this receiver should be built and tested one section at a time. Ugly construction easily allows this to be done. I started with the final amp and then worked backwards through the schematic until the antenna input was reached. Build the 2 low pass filters and the source follower as one section as the source follower is needed to bias the lowpass filter stages. The AF amp stages can be tested with a home brew AF oscillator such as a free-running multi vibrator.